Uplink channel estimation

ABSTRACT

In one embodiment, a receiver is provided for use in a multiple-input system that includes a receiving antenna receiving a time-domain signal corresponding to a plurality of signals transmitted from a plurality of transmitting antennas. The receiver includes: (a) a transform unit adapted to transform the time-domain signal into a frequency-domain signal; (b) a channel estimation unit adapted to estimate, based on the frequency-domain signal and a frequency-domain pilot signal, a combined transfer function corresponding to a plurality of transfer functions of respective channels between the plurality of transmitting antennas and the receiving antenna; and (c) a channel separation unit including a plurality of frequency-domain convolution units that separate the combined transfer function into a plurality of estimated channel transfer functions.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to channel estimation in a wirelesscommunications system, and, in particular, in a UMTS Long Term Evolution(LTE) system uplink.

2. Description of the Related Art

The Universal Mobile Telecommunications System (UMTS) is a high-speedcellular radio system that provides digital data and voicecommunications. UMTS has recently evolved from 3G systems to 3.5Gsystems using High-Speed Downlink Packet Access (HSDPA) and High-SpeedUplink Packet Access (HSUPA), and still continues to evolve. The UMTSLong Term Evolution (LTE) protocol is currently being specified in 3GPPRelease 8 to ensure its competitiveness for the next ten years andbeyond. LTE, which is also known as Evolved UMTS Terrestrial RadioAccess (UTRA) and Evolved UMTS Terrestrial Radio Access Network (UTRAN),provides new physical-layer concepts and protocol architectures forUMTS. See, e.g., Application Note 1MA111, “UMTS Long Term Evolution(LTE) Technology Introduction,” Rohde & Schwarz GmbH & Co. KG, availableathttp://www2.rohde-schwarz.com/en/service_and_support/Downloads/Application_Notes/,hereby incorporated by reference in its entirety.

According to the 3GPP Release 8 standard, the LTE downlink usesOrthogonal Frequency-Division Multiple Access (OFDMA). The LTE uplinkuses Single-Carrier FDMA (SC-FDMA), which allows a low-complexityreceiver implementation in the base station.

FIG. 1 depicts a simplified block diagram of an LTE uplink 100,including an LTE uplink transmitter 110 (e.g., located in a UMTS mobileterminal) and an LTE uplink receiver 130 (e.g., located in a UMTS basestation). In transmitter 110, serial data 112 to be transmitted isquadrature-amplitude-modulation (QAM) modulated by modulator 114 andconverted into N parallel streams by serial-to-parallel converter 116.The N parallel streams are then input to an inverse fast Fouriertransform (IFFT) unit 118, which uses the N parallel streams as bins tocreate a block of N OFDM symbols, each mapped to a different subcarrierfrequency. The N OFDM symbols output from IFFT unit 118 are serializedby parallel-to-serial converter 120 to create a time-domain OFDM signal,and cyclic extension unit 122 adds to the time-domain OFDM signal acyclic prefix that provides guard time to reduce intersymbolinterference. Finally, the time-domain OFDM signal is transmitted overthe LTE uplink.

In receiver 130, the transmitted signal is received and input to cyclicextension unit 132, which removes the cyclic extension. The resultingsignal is converted into N parallel OFDM symbols at serial-to-parallelconverter 134, and the N parallel OFDM symbols are input to fast Fouriertransform (FFT) unit 136, which removes the N subcarrier frequencies andoutputs N parallel words. The N parallel words are then input to (i) aparallel-to-serial converter 138, which converts the N parallel words toserial data, and (ii) a channel estimator unit 140, which outputs, foreach parallel word, an estimated channel transfer function, based on alocally generated pilot signal and a copy of the pilot signal containedwithin the received signal. M-point IDFT and equalizer unit 142 thenseparates the serial data for each channel via an M-point IDFT andequalizes the serial data for each channel, based on the correspondingestimated channel transfer function from channel estimator unit 140. Ina single-input, single-output (SISO) system having a single transmissionchannel from a single transmitter antenna to a single receiver antenna,channel estimator 140 generates an estimated channel transfer functioncorresponding to the single transmission channel. The equalized serialdata for each channel is then demodulated by QAM demodulator unit 144 torecover data 146.

To assist with channel estimation, the LTE uplink protocol includes thetransmission of known pilot signals at regular intervals, along withdata signals. As described above, channel estimator 140 in receiver 130uses these transmitted pilot signals to estimate channel characteristicsin the LTE uplink. M-point IDFT and equalizer unit 142 in receiver 130then uses the channel estimates to improve the accuracy of the datareception and demodulation. Conventional techniques for channelestimation, e.g., the Minimum Mean Square Error (MMSE) and Least Squares(LS) techniques, are computationally expensive, however.

An LTE uplink may also include advanced antenna technologies, such asMultiple Input Multiple Output (MIMO). See, e.g., A. Toskala et al.,“Utran Long Term Evolution in 3GPP,” IEEE 17th International Symposiumon Personal, Indoor and Mobile Radio Communications, pp. 1-5, September2006, hereby incorporated by reference in its entirety. In a MIMO-basedsystem, there are at least two transmitter antennas and at least tworeceiver antennas.

FIG. 2 represents a two-by-two MIMO LTE system 200 having twotransmitter antennas 202 and 204, which transmit signals over fourtransmission paths 206, 208, 210, and 212 to two receiver antennas 214and 216, where each receiver antenna 214, 216 receives a signalcorresponding to the superposition of signals arriving over twodifferent transmission paths. For example, the signal received atreceiver antenna 214 corresponds to the superposition of (i) the signaltransmitted from transmitter antenna 202 via transmission path 206 and(ii) the signal transmitted from transmitter antenna 204 viatransmission path 208. In general, each transmitter antenna 202, 204 isassociated with its own transmitter analogous to transmitter 110 of FIG.1, and each receiver antenna 214, 216 is associated with its ownreceiver analogous to receiver 130 of FIG. 1.

In order to separate the transmitted pilot signals from each MIMOtransmitter antenna 202, 204 received at each receiver antenna 214, 216,MIMO LTE system 200 may employ cyclic-shift transmit diversity (CSTD).CSTD is an adaptation of the idea of delay diversity to OFDM systems.With CSTD, each antenna element in a transmit array sends a circularlyshifted version of the same OFDM time-domain symbol. For eachtransmitter antenna, a cyclic prefix is added after circularly shiftingthe OFDM symbol. See, e.g., Javvin Technologies, Inc., WirelessTechnology Terms, Glossary and Dictionary, athttp://www.javvin.com/wireless/CSTD.html. For example,constant-amplitude zero-autocorrelation (CAZAC) sequences may beemployed to provide a CSTD transmission scheme. Thus, in MIMO LTE system200, the signal transmitted by transmitter antenna 204 is a circularlyshifted version of the signal transmitted by transmitter antenna 202.

The pilot signals received at each receiver antenna may be separated byconverting the OFDM signals into the time domain and performingwindowing operations based on the cyclic shift. For example, U.S. PatentApplication Publication No. US 2008/0031375 filed by Zhou et al.(hereinafter, “Zhou”), hereby incorporated by reference in its entirety,describes a channel estimation and separation method in a MIMO-OFDMsystem. The Zhou method includes (a) Fourier-transforming a plurality ofsignals received by a receiving antenna; (b) performing channelestimation by a least-squares method based on a known pilot signal; (c)inverse-Fourier-transforming the least-squares channel estimates into animpulse response in the time domain; (d) separating the transformedimpulse response into channel impulse responses of the respectivesignals by use of a window function having a low-pass filtercharacteristic; and finally (e) obtaining frequency-domain estimatedtransfer functions for each channel by Fourier-transforming each of thechannel impulse responses. The Zhou method, however, is computationallyexpensive and requires extensive processing resources.

SUMMARY OF THE INVENTION

In one embodiment, problems in the prior art are addressed by performingchannel estimation in the frequency domain using a matched filter, whichis computationally inexpensive. Further, channel separation is performedin the frequency domain, rather than in the time domain. By performingboth channel estimation and channel separation in the frequency domain,the Fourier transform and the inverse Fourier transform found in theprior art may be eliminated. As a result, a channel estimator inaccordance with one embodiment of the invention is simple and efficient.

Thus, in one embodiment, the invention is a receiver for use in amultiple-input system that includes a receiving antenna receiving atime-domain signal corresponding to a plurality of signals transmittedfrom a plurality of transmitting antennas. The receiver comprises: atransform unit adapted to transform the time-domain signal into afrequency-domain signal; a channel estimation unit adapted to estimate,based on the frequency-domain signal and a frequency-domain pilotsignal, a combined transfer function corresponding to a plurality oftransfer functions of respective channels between the plurality oftransmitting antennas and the receiving antenna; and a channelseparation unit including a plurality of frequency-domain convolutionunits that separate the combined transfer function into a plurality ofestimated channel transfer functions.

In another embodiment, the invention is a receiver for use in amultiple-input system that includes a receiving antenna receiving atime-domain signal corresponding to a plurality of signals transmittedfrom a plurality of transmitting antennas. The receiver comprises: atransform unit adapted to transform time-domain signal into afrequency-domain signal; a matched-filter unit adapted to estimate,based on the frequency-domain signal and the frequency-domain pilotsignal, a combined transfer function corresponding to a plurality oftransfer functions of respective channels between the plurality oftransmitting antennas and the receiving antenna; and a channelseparation unit adapted to separate the combined transfer function intoa plurality of estimated channel transfer functions.

In yet another embodiment, the invention is a method of estimating aplurality of estimated transfer functions corresponding to a pluralityof wireless channels in a multiple-input system that includes areceiving antenna receiving a time-domain signal corresponding to aplurality of signals transmitted from a plurality of transmittingantennas. The time-domain signal are transformed into a frequency-domainsignal. Based on the frequency-domain signal and a frequency-domainpilot signal, a combined transfer function corresponding to a pluralityof transfer functions of respective channels between the plurality oftransmitting antennas and the receiving antenna is estimated. Thecombined transfer function into a plurality of estimated channeltransfer functions are separated by convolving, in the frequency domain,the combined transfer function with a plurality of windowing functions.

In still another embodiment, the invention is a method of estimating aplurality of estimated transfer functions corresponding to a pluralityof wireless channels in a multiple-input system that includes areceiving antenna receiving a time-domain signal corresponding to aplurality of signals transmitted from a plurality of transmittingantennas. The time-domain signal is transformed into a frequency-domainsignal. A combined transfer function corresponding to a plurality oftransfer functions of respective channels between the plurality oftransmitting antennas and the receiving antenna is estimated byperforming a matched-filter operation on the frequency-domain signal andthe frequency-domain pilot signal; and the combined transfer function isseparated into a plurality of estimated channel transfer functions.

BRIEF DESCRIPTION OF THE DRAWINGS

Other aspects, features, and advantages of the present invention willbecome more fully apparent from the following detailed description, theappended claims, and the accompanying drawings, in which like referencenumerals identify similar or identical elements.

FIG. 1 is a block diagram of a prior-art LTE uplink, including atransmitter and a receiver.

FIG. 2 is a block diagram illustrating a MIMO system with twotransmitter antennas and two receiver antennas.

FIG. 3 is a timing diagram illustrating the structure of a radio framein an LTE uplink.

FIG. 4 is a block diagram of a channel estimator employing windowing inthe time domain in accordance with one embodiment of the invention.

FIG. 5 is a block diagram of a channel estimator employing windowing inthe time domain with fewer taps in accordance with another embodiment ofthe invention.

FIG. 6 is a block diagram of a channel estimator employing convolutionin the frequency domain in accordance with another embodiment of theinvention.

FIG. 7 is a table comparing the computational complexities of varioustime-domain estimation methods.

FIG. 8 is a table comparing the computational complexities of variousfrequency-domain estimation methods.

FIG. 9 is a graph illustrating the normalized error variance versus thenumber of taps for various signal-to-noise ratios (SNRs), evaluated at apilot signal block length of 72 subcarriers, in accordance with variousembodiments of the invention.

FIG. 10 is a graph illustrating the normalized error variance versus thenumber of taps for various SNRs, evaluated at a pilot signal blocklength of 576 subcarriers, in accordance with various embodiments of theinvention.

FIG. 11 is a table comparing the error variance with regard to variousnumbers of taps and various signal-to-noise ratios for varioustime-domain estimation methods.

FIG. 12 is a graph illustrating the normalized error variance versus thenumber of frequency components for various SNRs, evaluated at a pilotsignal block length of 72 subcarriers and a channel length of 7 taps inthe time domain, in accordance with various embodiments of theinvention.

FIG. 13 is a graph illustrating the normalized error variance versus thenumber of frequency components for different SNRs, evaluated at a pilotsignal block length of 576 subcarriers and a channel length of 72 tapsin the time domain, in accordance with various embodiments of theinvention.

FIG. 14 is a table comparing the error variance with regard to variousnumbers of taps and various signal-to-noise ratios for varioustime-domain and frequency-domain estimation methods.

FIGS. 15( a)-(b) are graphs illustrating the normalized error varianceversus SNR, when (a) the pilot signal block length is 72 subcarriers and(b) the pilot signal block length is 576 subcarriers, respectively, inaccordance with various embodiments of the invention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 3 shows the structure of an exemplary radio frame (known as a “type1” radio frame) in an LTE system. This frame structure is applicable toboth frequency-division duplex (FDD) and time-division duplex (TDD) LTEsystems. Each radio frame is T_(f)=307200×T_(s)=10 ms long and consistsof 20 slots of length T_(slot)=15360×T_(s)=0.5 ms, numbered from 0 to19. A sub-frame is defined as two consecutive slots. Depending on thecyclic prefix length, each slot consists of 6 or 7 OFDM symbols for anextended or normal cyclic prefix, respectively. Assuming that the systemuses a normal cyclic prefix, such that each slot has 7 OFDM symbols, thepilot signals transmitted for channel estimation occupy the fourth OFDMsymbol.

A received signal y is the sum of (i) the convolution of an input pilotsignal x with the channel impulse response h and (ii) Gaussian noise nas shown in Equation (1) below:

y=x*h+n   (1)

Equation (1) can be written in the frequency domain as shown in Equation(2) below:

{right arrow over (Y)}={right arrow over (X)}{right arrow over(H)}+{right arrow over (N)}  (2)

where {right arrow over (Y)},{right arrow over (X)},{right arrow over(H)}, and {right arrow over (N)} respectively represent the receivedsignal, transmitted signal, channel frequency response, and noise, allin the frequency domain. Equation (2) can be expressed as shown inEquation (3) below:

$\begin{matrix}{{\begin{bmatrix}y_{1} \\y_{2} \\\vdots \\y_{M}\end{bmatrix} = {{\begin{bmatrix}x_{1} & \; & \; & 0 \\\; & x_{2} & \; & \; \\\; & \; & ⋰ & \; \\0 & \; & \; & x_{M}\end{bmatrix}\begin{bmatrix}h_{1} \\h_{2} \\\vdots \\h_{M}\end{bmatrix}} + \begin{bmatrix}n_{1} \\n_{2} \\\vdots \\n_{M}\end{bmatrix}}},} & (3)\end{matrix}$

where the pilot signal x is placed along the diagonal elements of thematrix {right arrow over (X)}. See, e.g., L. Somasegaran, “ChannelEstimation and Prediction in UMTS LTE,” M.S. Thesis, Dept. of ElectronicSystems, Aalborg Univ., Aalborg, Denmark, 2007 (hereinafter,“Somasegaran”), hereby incorporated by reference in its entirety.Equation (2) can also be represented as shown in Equation (4) below:

{right arrow over (Y)}={right arrow over (X)}{right arrow over(H)}+{right arrow over (N)}  (4)

where {right arrow over (H)}={right arrow over (W)}{right arrow over(C)}, and {right arrow over (W)} is the discrete Fourier transform (DFT)matrix defined as

$W = {\frac{1}{\sqrt{M_{u}}}^{\frac{{- 2}j\; \pi \; i\; k}{M_{u}}}}$

for i=0, . . . ,M_(u)−1; k=0, . . . ,L−1, where M_(u) is the length ofthe pilot signal, and L is the number of taps. {right arrow over (C)} isthe channel impulse response with L taps.

FIG. 2 shows a multiple-input, multiple-output (MIMO) system with twotransmitter antennas 202, 204 and two receiver antennas 214, 216. Thereceived signal can be expressed as shown in Equation (5) below:

$\begin{matrix}{{{{\overset{->}{Y}}_{1} = {{\begin{bmatrix}{\overset{->}{X}}_{1} & {\overset{->}{X}}_{2}\end{bmatrix}\begin{bmatrix}{\overset{->}{H}}_{1} \\{\overset{->}{H}}_{2}\end{bmatrix}} + \begin{bmatrix}{\overset{->}{N}}_{1} \\{\overset{->}{N}}_{2}\end{bmatrix}}},{{\overset{->}{Y}}_{2} = {{\begin{bmatrix}{\overset{->}{X}}_{1} & {\overset{->}{X}}_{2}\end{bmatrix}\begin{bmatrix}{\overset{->}{H}}_{3} \\{\overset{->}{H}}_{4}\end{bmatrix}} + \begin{bmatrix}{\overset{->}{N}}_{3} \\{\overset{->}{N}}_{4}\end{bmatrix}}}}\text{where}{{{\overset{->}{Y}}_{1} = \begin{bmatrix}y_{11} & y_{12} & \ldots & y_{1M}\end{bmatrix}^{T}},{{\overset{->}{Y}}_{2} = \begin{bmatrix}y_{21} & y_{22} & \ldots & y_{2M}\end{bmatrix}^{T}}}} & (5)\end{matrix}$

${\begin{bmatrix}{\overset{->}{X}}_{1} & {\overset{->}{X}}_{2}\end{bmatrix} = \begin{bmatrix}x_{11} & \; & \; & 0 & x_{21} & \; & \; & 0 \\\; & x_{12} & \; & \; & \; & x_{22} & \; & \; \\\; & \; & ⋰ & \; & \; & \; & ⋰ & \; \\0 & \; & \; & x_{1M} & 0 & \; & \; & x_{2M}\end{bmatrix}},{\begin{bmatrix}{\overset{->}{H}}_{1} \\{\overset{->}{H}}_{2}\end{bmatrix} = \begin{bmatrix}h_{11} & h_{12} & \ldots & h_{1M} & h_{21} & h_{22} & \ldots & h_{2M}\end{bmatrix}^{T}},{\begin{bmatrix}{\overset{->}{H}}_{3} \\{\overset{->}{H}}_{4}\end{bmatrix} = \begin{bmatrix}h_{31} & h_{32} & \ldots & h_{3M} & h_{41} & h_{42} & \ldots & h_{4M}\end{bmatrix}^{T}},{\begin{bmatrix}{\overset{->}{N}}_{1} \\{\overset{->}{N}}_{2}\end{bmatrix} = \begin{bmatrix}n_{11} & n_{12} & \ldots & n_{1M} & n_{21} & n_{22} & \ldots & n_{2M}\end{bmatrix}^{T}},{\begin{bmatrix}{\overset{->}{N}}_{3} \\{\overset{->}{N}}_{4}\end{bmatrix} = {\begin{bmatrix}n_{31} & n_{32} & \ldots & n_{3M} & n_{41} & n_{42} & \ldots & n_{4M}\end{bmatrix}^{T}.}}$

In Equation (5), {right arrow over (X)}₁ and {right arrow over (X)}₂respectively represent the data transmitted from antennas 202 and 204 inFIG. 2, {right arrow over (Y)}₁ and {right arrow over (Y)}₂ respectivelyrepresent the data received at antennas 214 and 216 in FIG. 2, {rightarrow over (H)}₁,{right arrow over (H)}₂,{right arrow over (H)}₃, and{right arrow over (H)}₄ denote the channel frequency responses betweenthe transmitter and receiver antennas over channels 206, 208, 210, and212, respectively, and {right arrow over (N)}₁,{right arrow over(N)}₂,{right arrow over (N)}₃, and {right arrow over (N)}₄ denoteGaussian noise over those same channels. Equation (5) can further besimplified as shown in Equation (6) below:

$\begin{matrix}{\begin{matrix}{{\overset{->}{Y}}_{1} = {{\begin{bmatrix}{\overset{->}{X}}_{1} & {\overset{->}{X}}_{2}\end{bmatrix}\begin{bmatrix}{\overset{->}{W}{\overset{->}{C}}_{1}} \\{\overset{->}{W}{\overset{->}{C}}_{2}}\end{bmatrix}} + \begin{bmatrix}{\overset{->}{N}}_{1} \\{\overset{->}{N}}_{2}\end{bmatrix}}} \\{= {{{\begin{bmatrix}{\overset{->}{X}}_{1} & {\overset{->}{X}}_{2}\end{bmatrix}\begin{bmatrix}\overset{->}{W} & 0 \\0 & \overset{->}{W}\end{bmatrix}}\begin{bmatrix}{\overset{->}{C}}_{1} \\{\overset{->}{C}}_{2}\end{bmatrix}} + \begin{bmatrix}{\overset{->}{N}}_{1} \\{\overset{->}{N}}_{2}\end{bmatrix}}} \\{= {{\overset{->}{E}{\overset{->}{F}\begin{bmatrix}{\overset{->}{C}}_{1} \\{\overset{->}{C}}_{2}\end{bmatrix}}} + \begin{bmatrix}{\overset{->}{N}}_{1} \\{\overset{->}{N}}_{2}\end{bmatrix}}}\end{matrix}{{\overset{->}{Y}}_{2} = {{\overset{->}{E}{\overset{->}{F}\begin{bmatrix}{\overset{->}{C}}_{3} \\{\overset{->}{C}}_{4}\end{bmatrix}}} + \begin{bmatrix}{\overset{->}{N}}_{3} \\{\overset{->}{N}}_{4}\end{bmatrix}}}} & (6) \\{{\text{where}\mspace{14mu} \overset{->}{E}} = {{\begin{bmatrix}{\overset{->}{X}}_{1} & {\overset{->}{X}}_{2}\end{bmatrix}\mspace{14mu} {and}\mspace{14mu} \overset{->}{F}} = \begin{bmatrix}\overset{->}{W} & 0 \\0 & \overset{->}{W}\end{bmatrix}}} & (7)\end{matrix}$

In Equations (6) and (7), {right arrow over (W)} denotes the DFT matrix,and {right arrow over (C)}₁,{right arrow over (C)}₂,{right arrow over(C)}₃, and {right arrow over (C)}₄ denote the impulse responses of thechannels between the transmitter and receiver antennas. {right arrowover (E)} denotes the transmitted data, and {right arrow over (F)}denotes the DFT matrices of all the transmitters.

Equation (6) will have a unique solution if and only if

${L \leq \frac{M_{u}}{T_{N}}},$

where T_(N) denotes the number of transmitters, M_(u) is the length ofthe pilot signal, and L is the number of taps. This problem has beensolved using least-squares and minimum mean square error methods invarious publications, including: (a) Somasegaran; (b) A. Ancora et al.,“Down-Samples Impulse Response Least-Squares Channel Estimation for LTEOFDMA,” IEEE International conference on Acoustics, Speech and SignalProcessing, vol. 3, pp. 293-296, April 2007 (hereinafter, “Ancora”); (c)J. V. D. Beek et al., “On Channel Estimation in OFDM Systems,” InProceeding of Vehicular Technology Conference, vol. 2, pp. 815-819,September 1995 (hereinafter, “Beek”); (d) U.S. Patent Application No. US2007/0014272 A1, entitled “Pilot and Data Transmission in aQuasi-Orthogonal Single-Carrier Frequency Division Multiple AccessSystem,” by R. Palanki et al., published Jan. 18, 2007 (hereinafter,“Palanki”); and (e) K. Eriksson, “Channel Tracking versus FrequencyHopping for Uplink LTE.” M.S. project, KTH School of ElectricalEngineering, Stockholm, Sweden, 2007 (hereinafter, “Eriksson”). Ancora,Beek, Palanki, and Eriksson are hereby incorporated by reference intheir entirety.

In particular, for a channel estimator employing MMSE estimation in thetime domain, and assuming that the channel vector {right arrow over (C)}is Gaussian and independent of channel noise N, the MMSE estimate

is given by Equation (8) below:

$\begin{matrix}{{{\overset{\overset{->}{\hat{}}}{C} = {{\overset{->}{R}}_{cy}{\overset{->}{R}}_{yy}^{- 1}\overset{->}{Y}}},\mspace{14mu} \text{where}}{{{\overset{->}{R}}_{cy} = {{E\left\{ {\overset{->}{C}{\overset{->}{Y}}^{H}} \right\}} = {{\overset{->}{R}}_{cc}{\overset{->}{F}}^{H}{\overset{->}{E}}^{H}}}},\mspace{14mu} {and}}{{{\overset{->}{R}}_{yy} = {{E\left\{ {\overset{->}{Y}{\overset{->}{Y}}^{H}} \right\}} = {{\overset{->}{E}\overset{->}{F}{\overset{->}{R}}_{cc}{\overset{->}{F}}^{H}{\overset{->}{E}}^{H}} + {\sigma_{n}^{2}{\overset{->}{I}}_{M}}}}},}} & (8)\end{matrix}$

where {right arrow over (Y)} is the received signal, {right arrow over(R)}_(cy) is the cross-covariance matrix between the channel vector{right arrow over (C)} and the received signal {right arrow over (Y)},and {right arrow over (R)}_(yy) is the auto-covariance matrix of thereceived signal {right arrow over (Y)}. {right arrow over (R)}_(cc) isthe auto-covariance matrix of the channel vector {right arrow over (C)},σ_(n) ² denotes the noise variance, {right arrow over (E)} and {rightarrow over (F)} respectively denote the transmitted data and the DFTmatrices of all the transmitters, as defined in Equation (7) above, Hdenotes the Hermitian conjugate operator, and {right arrow over (I)}_(M)is the identity matrix. See, e.g., Beek.

Further, for a channel estimator employing LS estimation in the timedomain, the LS estimator Ĉ for impulse response can be expressed asshown in Equation (9):

$\begin{matrix}\begin{matrix}{\hat{C} = {\left( {({EF})^{H}({EF})} \right)^{- 1}({EF})^{H}\overset{->}{Y}}} \\{= {\left( {F^{H}E^{H}{EF}} \right)^{- 1}F^{H}E^{H}\overset{->}{Y}}} \\{= {M_{u}F^{H}E^{H}\overset{->}{Y}}}\end{matrix} & (9)\end{matrix}$

where {right arrow over (Y)} is the received signal, {right arrow over(E)} and {right arrow over (F)}, respectively, denote the transmitteddata and the DFT matrices of all the transmitters, as defined inEquation (7) above, H denotes the Hermitian conjugate operator, andM_(u) is the length of the pilot signal. See, e.g., Palanki.

In certain embodiments of the present invention, uplink channels in aMIMO LTE system employing cyclic-shift transmit diversity (CSTD) areestimated using various new methods for channel estimation in the timeand frequency domains.

FIG. 4 shows a block diagram of channel estimator 400, which performs awindowing method (WM) for channel estimation, according to oneembodiment of the present invention. Channel estimator 400 may be usedto implement channel estimator 140 in receiver 130 of FIG. 1. Channelestimator 400 includes an estimation unit 402 and a channel separationunit 406. Channel separation unit 406 includes an M-point inversediscrete Fourier transform (IDFT) unit 408, T_(N) windowing units 412,and T_(N) corresponding M-point (DFT) units 418, wherein T_(N)corresponds to the number of transmitter antennas in the MIMO system.

Estimation unit 402 is preferably implemented as a frequency-domainmatched filter, as shown in FIG. 4. Generally, a matched filter is afilter that correlates a known signal with an unknown signal,conventionally to detect the presence of the known signal in the unknownsignal. In the time domain, a matched filter is equivalent to convolvingthe unknown signal with a time-reversed version of the template (e.g.,cross-correlation). A frequency-domain matched filter may be implementedby performing bin-by-bin multiplication of the frequency-domainrepresentations of the known signal and the unknown signal.Alternatively, estimation unit 402 may be implemented using MMSE or LStechniques.

Estimation unit 402 receives, at its input, the N parallel wordsoutputted by FFT 136 shown in FIG. 1, as well as a locally generatedfrequency-domain representation of the pilot signal. Assuming thatestimation unit 402 is implemented as a frequency-domain matched filteras shown in FIG. 4, estimation unit 402 correlates these two signals inthe frequency domain by multiplying the frequency-domain received signalby the frequency-domain pilot signal (or vice versa). The result is afrequency-domain correlation signal 404 having M frequency components.Correlation signal 404 represents the combined transfer functioncorresponding to the transfer functions of the channels between thetransmitting antennas (e.g., antennas 202 and 216 of FIG. 2) and thecorresponding receiver antenna (e.g., either antenna 214 or antenna 216of FIG. 2).

Correlation signal 404 is then input to M-point IDFT unit 408, whichperforms an M-point IDFT to convert correlation signal 404 into atime-domain signal 410. Time-domain signal 410 consists of theconcatenation of the impulse responses from all of the transmitterantennas to the receiver antenna, separated by a time delay t of onecyclic shift, assuming that the transmitted signals are formatted inaccordance with a cyclic-shift transmit diversity (CSTD) scheme.

Time-domain signal 410 is then input to T_(N) windowing units 412, inwhich windowing operations are performed on time-domain signal 410 atT_(N) different delay values t_(k) (where k=1 . . . T_(N)), using awindow of the size of the channel impulse response duration at thesignal bandwidth rate M_(u)/T_(N). Delay values t_(k) are preferablydefined by t_(k)=k(cyclic shift period), where k=1 . . . T_(N), andT_(N) is the number of transmitter antennas 202, 204. The channelimpulse response duration is predetermined as a system parameter duringthe design of transmitter antennas 202, 204. The precise location intime of each window is preferably chosen such that the maximum peak islocated in the center of the window.

Finally, the windowed, time-domain signal 414 outputted from eachwindowing unit 412 is input to a corresponding M-point DFT unit 418,which performs an M-point DFT on signal 414 and produces a correspondingfrequency-domain output signal 420. Output signals 420 represent thefrequency-domain channel responses of the T_(N) channels, whichresponses may then be used as channel estimates.

FIG. 5 shows a block diagram of channel estimator 500, which performs awindowing method with fewer taps (WMFT) for channel estimation,according to another embodiment of the present invention. Channelestimator 500 is similar to channel estimator 400 of FIG. 4 with similarelements and signals identified using labels having the same last twodigits, except that separation unit 506 includes T_(N) thresholdingunits 516 inserted between windowing units 512 and M-point DFT units518. Thresholding units 516 compare the power of each tap (i.e.,discrete sample) of the windowed, time-domain signals 514 with a noisethreshold, and pass to the M-point DFT units 518 only those taps havinga power greater than the noise threshold. Taps within signals 514 havinga power lower than the noise threshold are dropped or zeroed out. Inthis manner, fewer than M_(u)/T_(N) channel taps are selected forM-point DFT conversion by M-point DFT units 518. Because only those tapshaving a power greater than the noise floor are used for channelestimation, the resulting channel estimates 520 are more accurate.Moreover, the processing overhead associated with performing M-pointDFTs for taps having a power less than the noise floor is avoided. Thistechnique of reducing the channel taps from M_(u)/T_(N) channel taps toL channel taps (where L is less than M_(u)/T_(N)) may be employed forany estimation technique, including estimation via MMSE, LS, andmatched-filtering.

FIG. 6 shows a block diagram of channel estimator 600, which performs afrequency-domain method (FDM) for channel estimation, according to yetanother embodiment of the present invention. Channel estimator 600comprises a frequency-domain matched filter 602 and channel separationunit 606 having T_(N) frequency-domain convolution units 622.Frequency-domain matched filter 602 receives, at its input, the Nparallel words outputted by FFT 136 shown in FIG. 1, as well as alocally generated frequency-domain representation of the pilot signal.Frequency-domain matched filter 602 correlates, in the frequency domain,these two signals by multiplying the frequency-domain received signal bythe conjugate of the frequency-domain pilot signal (or vice versa). Theresult is a frequency-domain correlation signal 604 having M frequencycomponents.

In contrast to the embodiments of FIGS. 4 and 5, however,frequency-domain correlation signal 604 is then input to T_(N)frequency-domain convolution units 622, which circularly convolve, inthe frequency domain, correlation signal 604 with T_(N) sinc signals(defined as sin(πx)/πx), each having a different phase shift. (Acircular convolution of two functions is defined in terms of theperiodic extension of one or both functions. Periodic extension means anew function is formed by shifting the original function by multiples ofa period T and adding all the copies together. See, e.g.,http://en.wikipedia.org/wiki/Circular_convolution, hereby incorporatedby reference in its entirety.) Because a sinc function in the frequencydomain corresponds to a windowing function in the time domain, theoutputs from frequency-domain convolution units 622 constitute thechannel frequency responses, which may then be used as channelestimates.

In another embodiment, which may be referred to as the “Frequency-DomainMethod with Few Frequency Components” (FDMFFC) for channel estimation,truncated sinc signals, rather than full sinc signals, are convolvedwith correlation signal 604 at convolution units 622. Whereas a fullsinc signal has a number of lobes approaching infinity, the truncatedsinc signals have a predetermined number of lobes, which may berepresented by a relatively small number of frequency components, e.g.,preferably between about 5 and about 20 components, and more preferablyabout 11 components. This reduction in the number of frequencycomponents of the sinc signals yields saving in the computationalcomplexity of the channel estimator while preserving the estimatorperformance.

In still another embodiment, which may be referred to as the “ModifiedFrequency-Domain Method with Few Frequency Components” (MFDMFFC) forchannel estimation, frequency-domain convolution units 622 perform (i)normal or linear convolution on the frequency components of correlationsignal 604 that correspond to end subcarriers of the received LTE uplinksignal and (ii) circular convolution on the frequency components thatcorrespond to the remaining middle subcarriers. End subcarriers are thesubcarriers at or near the lowest and highest subcarrier frequencies inthe pilot signal, while middle subcarriers are the remainingsubcarriers. In one embodiment, about 6 subcarriers in the lowestfrequencies and about 5 subcarriers in the highest frequencies may besubjected to linear convolution, and the remaining subcarriers subjectedto circular convolution). Performing linear convolution on the endsubcarriers' frequency components mitigates edge leakage (i.e., theunintended influence that the highest frequencies and lowest frequencieshave on each other in circular convolution, even though they areuncorrelated in practice) that otherwise would arise, because the cyclicapplication of the sinc function averages across the two edges (i.e.,the highest and lowest subcarrier frequencies.)

Complexity Analysis

FIGS. 6 and 7 respectively show Tables 1 and 2. Table 1 indicatescomputational complexities in terms of number of operations required forchannel estimation using the time-domain methods of WM with estimationby MMSE with tap reduction (WMMMSE), by LS with tap reduction (WMLS), bymatched-filtering without tap reduction (WMMF), and by matched-filteringwith tap reduction (WMFT), for each receiver antenna. Table 2 indicatescomputational complexities for channel estimation using thefrequency-domain methods of FDM, FDMFFC, and MFDMFFC for each receiverantenna.

In all seven methods, received signal Rx_Data is in the time domain, andan M-point DFT is performed to convert the received signal into thefrequency domain. See, e.g., row 702 of Table 1. This M-point DFToperation requires O(M log M) arithmetic operations.

The computation of the inverse of the matrix {right arrow over (R)}_(yy)for the MMSE estimation of Equation (8), shown in row 704 of Table 1,requires O(M³) arithmetic operations.

To perform the estimation calculation, e.g., at block 402 of FIG. 4 andat block 502 of FIG. 5, various numbers of arithmetic operations arerequired for each time-domain method, as shown in row 706 of Table 1 inFIG. 7. In particular, WMMMSE and WMLS require T_(N) O(M) operations,while WMMF and WMFT require O(M) operations.

In all the time-domain methods, the estimated signal (e.g., signal 404in FIG. 4 and signal 504 in FIG. 5) is converted back to the time domainby performing an M-point IDFT operation (e.g., at block 408 of FIG. 4and at block 508 of FIG. 5). In this process, as shown in row 708 ofTable 1, WMMMSE, WMLS, and WMFT all require O(ML) arithmetic operations,while WMMF requires O(M log M) arithmetic operations (because no tapreduction is performed in the WMMF method.)

Finally, for all four time-domain methods, converting each estimatedchannel into the frequency domain to produce an estimated channelfrequency response requires T_(N) O(M log M) arithmetic operations, asshown in row 710 of Table 1.

Among all the time-domain methods, assuming that the down-sampledchannel order L is less than log M, WMFT has the least complexity, whileWMMMSE has the highest. Unfortunately, however, the number of tapsrequired in the WMFT method varies with the length M_(u) of the pilotsignal and the signal-to-noise ratio (SNR).

Among the frequency-domain methods, from Table 2, it may be observedthat FDMFFC and MFDMFFC have lesser complexities than FDM, assuming thatL is less than log M. By comparing FIGS. 9 and 10, it may be seen that,unlike WMFT, the number of frequency components needed in FDMFFC (and,therefore, in MFDMFFC as well) is not a function of the pilot-signallength M_(u). This difference makes the complexity of MFDMFFC less thanthat of WMFT, especially for long pilot signals. Moreover, this savingin complexity comes with either no degradation or actual improvement inthe accuracy of the channel estimation.

Simulation Results

The methods discussed above were simulated in a MATLAB® tool by TheMathWorks, Inc., of Natick, Mass., using the pseudocode attached in theAppendix and the following parameters. The channel impulse response withI_(C) taps was defined for the entire bandwidth. Depending on the lengthM_(u) of the pilot signal, the channel tap estimates can be down-sampledfrom the channel order I_(C) to a preferred channel order L, preferablyselected to obtain the lowest possible error variance. The channel orderI_(C) can be defined in terms of transmitted bandwidth B and coherencebandwidth B_(C) according to the following equation: I_(C)=B/B_(C). Thecoherence bandwidth is defined as

${B_{c} \approx \frac{1}{5\; \tau}},$

where τ is the multi-path time-delay spread. In an LTE system, themaximum cyclic prefix is 160 samples, which, in the time domain, isequal to 5.2 μs (given by 160/f_(s), where f_(s) is the samplingfrequency, equal to 30.72 MHz). Generally, the cyclic prefix is usuallychosen to be 4-5 times the worst-case delay spread. Hence, the biggestdelay spread that this simulated embodiment of the system can mitigateis 1.3 μs (5.2/4). Consequently, the coherence bandwidth B_(C) is 154kHz. For a transmitted bandwidth of 20 MHz, the channel order I_(C) is133.

The normalized error variance can be computed as

-{right arrow over (C)}|², where

and {right arrow over (C)} are the estimated and actual channelfrequency responses, respectively. The performance of the error variancewith regard to various numbers of taps and various SNRs, usingtime-domain methods when four transmitters transmit the signal, areshown in FIGS. 9 and 10 for block lengths of 72 and 576 subcarriers,respectively. The results are tabulated in Table 3 in FIG. 11. FromFIGS. 9 and 10, it can be concluded that, after reaching the lowesterror variance, the error performance for a low SNR decreases with anincrease in the order of the channel. This decrease in error performancearises from the addition of estimates with a poor SNR. However, for ahigh SNR, the error performance remains the same or has a negligibledifference. From FIGS. 9 and 10, for block lengths of 72 and 576subcarriers, WMMF with 7 and 70 taps, respectively, gives a betterperformance when compared to other methods.

The error performance of WMMF and WMFT for a block length of 72subcarriers with 18 and 7 taps, respectively, as compared with FDM andFDMFFC, is shown in FIG. 12. For the block length of 576 subcarriers,FIG. 13 compares the error performance of FDM and FDMFFC with the errorperformance of WMMF and WMFT with 144 and 70 taps, respectively. It canbe observed that about 11 frequency components are required to representthe sinc signal, in order to have the error performance comparable orbetter than that of WMFT. These results are tabulated in Table 4 in FIG.14. Depending on overall complexity and error performance, one of theabove methods can be selected for a given application.

The complexity of WMFT when the length M_(u) of the pilot signal is 72and 576 subcarriers (with number of taps equal to 7 and 70,respectively) is 3816 and 75456, respectively (using a mixed-radiximplementation for IDFT unit 508 and DFT unit 518 shown in FIG. 5).Similarly, for FDMFFC, the complexity for length M_(u) equal to 72 and576 is 3888 and 32832, respectively, when 11 frequency components arechosen for both cases. For a small length M_(u), the complexity of WMFTand FDMFFC is comparable, whereas, for large length M_(u), FDMFFC hasless complexity than WMFT. In the case of length M_(u) equal to 576subcarriers, there is a complexity savings of 56.5%, when FDMFFC is usedinstead of WMMF.

The error performance with regard to the SNR for FDMFFC and MFDMFFC,when block lengths of 72 and 576 are used, is shown in FIGS. 15( a) and15(b). In FIGS. 15( a)-(b), FDMFFC and MFDMFFC represent the errorperformance when all the subcarriers of the pilot signal block areconsidered. “Mid SC” and “End SC” represent the middle subcarriers(e.g., subcarriers M-11) and end subcarriers (e.g., in one embodiment,about 6 subcarriers in the beginning and about 5 subcarriers at the endof the pilot signal block of length M_(u)) that are considered for errorcomputation. From FIGS. 15( a)-(b), it can be observed that asignificant amount of error is introduced by the end subcarriers.Performing linear convolution on the end subcarriers in MFDMFFCmitigates the error in the end subcarriers when compared with circularconvolution in FDMFFC.

Based on the above discussion and results, the following conclusions canbe surmised. Among all the time-domain methods, WMFT has the bestperformance, when complexity is considered. With an increase in thepilot signal block length M_(u), the number of taps required in WMFTincreases, whereas the number of frequency components required forrepresenting the sinc function in the frequency-domain methods remainsconstant. This constancy of the number of frequency components in thefrequency-domain methods considerably simplifies the implementation ofthe frequency-domain methods over time-domain methods and results in asignificant saving in computation complexity for large pilot signalblock sizes. Further, MFDMFFC advantageously mitigates the degradationof end subcarriers caused by edge leakage.

There has thus been described a number of embodiments of a channelestimator that provide better performance, with less complexity, thanexisting channel estimators. It will be understood, however, thatvarious changes in the details, materials, and arrangements of the partswhich have been described and illustrated in order to explain the natureof this invention may be made by those skilled in the art withoutdeparting from the scope of the invention as expressed in the followingclaims. For example, while certain embodiments above are described asemploying convolution of a matched filter output signal with a sincsignal—which is a frequency-domain representation of a rectangularwindow function in the time domain—frequency-domain representations ofother windowing functions may also be used in place of the sinc signal.Such windowing functions may include, but are not limited to, a Hammingwindow, a Hann window, a cosine window, a Lanczos window, a Bartlettwindow, a triangular window, a Gaussian window, a Bartlett-Hann window,a Blackman window, a Kaiser window, a Nuttall window, a Blackman-Harriswindow, a Blackman-Nuttall window, a flat-top window, a Bessel window,an exponential window, and a Tukey window.

It should further be understood that, although channel estimation in theabove embodiments is preferably provided by a matched filter, otherchannel estimation techniques, such as MMSE and LS techniques, may beused in accordance with the present invention.

The present invention may be implemented as analog, digital, or a hybridof both analog and digital circuit based processes, including possibleimplementation as a single integrated circuit (such as an ASIC or anFPGA), a multi-chip module, a single card, or a multi-card circuit pack.As would be apparent to one skilled in the art, various functions ofcircuit elements may also be implemented as processing blocks in asoftware program. Such software may be employed in, for example, adigital signal processor, micro controller, or general purpose computer.

Also for purposes of this description, the terms “couple,” “coupling,”“coupled,” “connect,” “connecting,” or “connected” refer to any mannerknown in the art or later developed in which energy is allowed to betransferred between two or more elements, and the interposition of oneor more additional elements is contemplated, although not required.Conversely, the terms “directly coupled,” “directly connected,” etc.,imply the absence of such additional elements.

Signals and corresponding nodes or ports may be referred to by the samename and are interchangeable for purposes here.

The present invention can be embodied in the form of methods andapparatuses for practicing those methods. The present invention can alsobe embodied in the form of program code embodied in tangible media, suchas magnetic recording media, optical recording media, solid statememory, floppy diskettes, CD-ROMs, hard drives, or any othermachine-readable storage medium, wherein, when the program code isloaded into and executed by a machine, such as a computer, the machinebecomes an apparatus for practicing the invention. The present inventioncan also be embodied in the form of program code, for example, whetherstored in a storage medium, loaded into and/or executed by a machine, ortransmitted over some transmission medium or carrier, such as overelectrical wiring or cabling, through fiber optics, or viaelectromagnetic radiation, wherein, when the program code is loaded intoand executed by a machine, such as a computer, the machine becomes anapparatus for practicing the invention. When implemented on ageneral-purpose processor, the program code segments combine with theprocessor to provide a unique device that operates analogously tospecific logic circuits.

Unless explicitly stated otherwise, each numerical value and rangeshould be interpreted as being approximate as if the word “about” or“approximately” preceded the value of the value or range.

The use of figure numbers and/or figure reference labels in the claimsis intended to identify one or more possible embodiments of the claimedsubject matter in order to facilitate the interpretation of the claims.Such use is not to be construed as necessarily limiting the scope ofthose claims to the embodiments shown in the corresponding figures.

It should be understood that the steps of the exemplary methods setforth herein are not necessarily required to be performed in the orderdescribed, and the order of the steps of such methods should beunderstood to be merely exemplary. Likewise, additional steps may beincluded in such methods, and certain steps may be omitted or combined,in methods consistent with various embodiments of the present invention.

Although the elements in the following method claims, if any, arerecited in a particular sequence with corresponding labeling, unless theclaim recitations otherwise imply a particular sequence for implementingsome or all of those elements, those elements are not necessarilyintended to be limited to being implemented in that particular sequence.

Reference herein to “one embodiment” or “an embodiment” means that aparticular feature, structure, or characteristic described in connectionwith the embodiment can be included in at least one embodiment of theinvention. The appearances of the phrase “in one embodiment” in variousplaces in the specification are not necessarily all referring to thesame embodiment, nor are separate or alternative embodiments necessarilymutually exclusive of other embodiments. The same applies to the term“implementation.”

1. A receiver for use in a multiple-input system that includes areceiving antenna receiving a time-domain signal corresponding to aplurality of signals transmitted from a plurality of transmittingantennas, the receiver comprising: a transform unit (e.g., 136) adaptedto transform the time-domain signal into a frequency-domain signal; achannel estimation unit (e.g., 602) adapted to estimate, based on thefrequency-domain signal and a frequency-domain pilot signal, a combinedtransfer function corresponding to a plurality of transfer functions ofrespective channels between the plurality of transmitting antennas andthe receiving antenna; and a channel separation unit (e.g., 606)including a plurality of frequency-domain convolution units (e.g., 622)that separate the combined transfer function into a plurality ofestimated channel transfer functions.
 2. The receiver of claim 1,wherein each convolution unit convolves, in the frequency domain, thecombined transfer function with a respective frequency-domainrepresentation of a windowing function.
 3. The receiver of claim 2,wherein the windowing function for each convolution unit isphase-shifted from the windowing function of each other convolutionunit.
 4. The receiver of claim 2, wherein each convolution unitcircularly convolves, in the frequency domain, the combined transferfunction with the respective frequency-domain representation of thewindowing function.
 5. The receiver of claim 2, wherein: the combinedtransfer function includes components corresponding to end subcarriersand components corresponding to middle subcarriers, and each convolutionunit (a) circularly convolves, in the frequency domain, the componentscorresponding to the middle subcarriers with the respectivefrequency-domain representation of the windowing function, and (b)linearly convolves, in the frequency domain, the componentscorresponding to the end subcarriers with the respectivefrequency-domain representation of the windowing function.
 6. Thereceiver of claim 5, wherein linearly convolving the componentscorresponding to the end subcarriers with the respectivefrequency-domain representation of the windowing function reduces edgeleakage between higher-frequency end subcarriers and lower-frequency endsubcarriers.
 7. The receiver of claim 2, wherein the frequency-domainrepresentation of the windowing function is a sinc signal.
 8. Thereceiver of claim 2, wherein the frequency-domain representation of thewindowing function is frequency-truncated.
 9. The receiver of claim 8,wherein the frequency-truncated, frequency-domain representation of thewindowing function is a sinc signal having a predetermined number offrequency components.
 10. The receiver of claim 1, wherein the channelestimation unit comprises a matched filter unit (e.g., 402, 502, 602)adapted to estimate, based on the frequency-domain signal and thefrequency-domain pilot signal, the combined transfer function.
 11. Thereceiver of claim 10, wherein the matched-filter unit estimates thecombined transfer function by performing bin-by-bin multiplication ofthe frequency-domain signal and the frequency-domain pilot signal. 12.The receiver of claim 1, wherein the channel estimation unit comprisesone of a minimum mean-square-error unit and a least-square unit.
 13. Areceiver for use in a multiple-input system that includes a receivingantenna receiving a time-domain signal corresponding to a plurality ofsignals transmitted from a plurality of transmitting antennas, thereceiver comprising: a transform unit (e.g., 136) adapted to transformtime-domain signal into a frequency-domain signal; a matched-filter unit(e.g., 402, 502, 602) adapted to estimate, based on the frequency-domainsignal and the frequency-domain pilot signal, a combined transferfunction corresponding to a plurality of transfer functions ofrespective channels between the plurality of transmitting antennas andthe receiving antenna; and a channel separation unit (e.g., 406, 506,606) adapted to separate the combined transfer function into a pluralityof estimated channel transfer functions.
 14. The receiver of claim 13,wherein the channel separation unit (e.g., 406, 506) comprises: aninverse transform unit (e.g., 408, 508) adapted to convert the combinedtransfer function into a time-domain combined impulse response signal; aplurality of windowing units (e.g., 412, 512), each adapted to separatethe time-domain combined impulse response signal into a correspondingchannel impulse response in the time domain, based on a correspondingphase-shifted windowing function; and a plurality of second transformunits (e.g., 418, 518), each adapted to convert the correspondingchannel impulse response into a frequency-domain estimate of thetransfer function of a corresponding channel.
 15. The receiver of claim14, wherein the length of the window of each phase-shifted windowingfunction is substantially equal to the channel impulse responseduration.
 16. The receiver of claim 14, wherein: the channel separationunit further comprises a plurality of thresholding units (e.g., 516),each connected between a corresponding windowing unit and acorresponding second transform unit; and each thresholding unit isadapted: (a) to determine a signal power of the respective channelimpulse response, (b) to compare the signal power with a thresholdvalue, and (c) to pass the respective channel impulse response to thecorresponding second transform unit only if the signal power of therespective channel impulse response is greater than a predeterminedlevel.
 17. The receiver of claim 16, wherein the predetermined level isthe signal-to-noise ratio of the plurality of received signals.
 18. Amethod of estimating a plurality of estimated transfer functionscorresponding to a plurality of wireless channels in a multiple-inputsystem that includes a receiving antenna receiving a time-domain signalcorresponding to a plurality of signals transmitted from a plurality oftransmitting antennas, the method comprising: transforming thetime-domain signal into a frequency-domain signal; estimating, based onthe frequency-domain signal and a frequency-domain pilot signal, acombined transfer function corresponding to a plurality of transferfunctions of respective channels between the plurality of transmittingantennas and the receiving antenna; and separating the combined transferfunction into a plurality of estimated channel transfer functions byconvolving, in the frequency domain, the combined transfer function witha plurality of windowing functions.
 19. A method of estimating aplurality of estimated transfer functions corresponding to a pluralityof wireless channels in a multiple-input system that includes areceiving antenna receiving a time-domain signal corresponding to aplurality of signals transmitted from a plurality of transmittingantennas, the method comprising: transforming the time-domain signalinto a frequency-domain signal, estimating a combined transfer functioncorresponding to a plurality of transfer functions of respectivechannels between the plurality of transmitting antennas and thereceiving antenna, by performing a matched-filter operation on thefrequency-domain signal and the frequency-domain pilot signal; andseparating the combined transfer function into a plurality of estimatedchannel transfer functions.